Lectures: MW in Urbauer Textbook: Dally and Poulton, Digital Systems Engineering. Grading: Approximate weighting for grade determination. The book introduces the topic of digital systems engineering by describing the major engineering problems associated with digital systems and the technology. 𝗣𝗗𝗙 | What makes some computers slow? What makes some digital systems operate reliably for years while others fail mysteriously every few.
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1 - INTRODUCTION TO DIGITAL SYSTEMS ENGINEERING. pp · https://doi. org//CBO Access. PDF; Export citation. See the course policy sheet for details. Lectures: MW to in Skilling Textbook: Dally and Poulton, Digital Systems Engineering. Skim Dally/Poulton “Digital Systems Engineering”. Chapter 3. Skim Overview Paper: bestthing.info; Includes running Stat Eye.
As described in Section 7.
In this case a parallel termination at the far end of the line is required to achieve incident-wave switching. This is not possible in a purely source-terminated configuration because. Unfortunately the amount of current sourced by a saturated FET can vary by 2: Several time constants are required for the output current to settle within a small percentage of kI ref.
In this configuration an inverter with V as its supply voltage is used to drive the gate of the current-source bias device.
With this input-gated configuration the output device can be made about half the size of each of the two series devices in the output-switched driver without running out of headroom across process comers.
The series connection can be eliminated by gating the bias voltage at the current source. With the switched driver. In both configurations. This small size is not without cost. When input. I ref. Placing two devices in series requires that the devices be made large and Vbias be made small to provide enough headroom on the current source to allow for the voltage drop across the resistive switching device.
This results in a driver with one-quarter the area and one-half the output capacitance of the switched driver. When in is high the driver sinks current kIref from the line. When in is low. With the gated driver. Vg is driven to ground. V is applied to the gate of a scaled PET. When in is high.
The gated driver has the disadvantage of a slow transient response compared with the output switched driver. Figure bias b. V ' In the first configuration. When in switches. The signal return in tum is AC-shorted to the termination supply. The predriver converts the full-swing differential inputs.
The other end of the line is parallelterminated into a positive termination voltage. With gain less than unity.. The loads of the predriver are set so that the swing of the gate drive signals is limited to the amount needed to fully switch the differential output pair.
The current-steering driver is naturally suited to drive a balanced differential line. This driver gets its speed by virtue of its fractional stage gain full-swing to limited-swing in the predriver. The complementary outputs of the driver are attached to the two conductors of the line. To reduce output delay. The differential driver is also effective in driving a single-ended line. As we noted in Section 4. In practice.. There is no dead band at the beginning or end of the signal swing This gives minimum output delay.
Owing to the shared signal return. The net result again is that the termination supply sees only DC current because the sum of the signal and return current for each driver is a constant. A bipolar driver can be constructed from any of the unipolar drivers described in the previous section by connecting two complementary unipolar drivers together. This requires that.
When the input. In the opposite state. With a unipolar driver. The received voltage. The receiver can discriminate between these two states by comparing the received current to zero.
VNbias and VPbias. A bipolar current-mode driver both sources and sinks current from the line. The output stage of this circuit is a pair of complementary current sources switched by a pair of complementary switch devices. Figure b shows the specific example of a bipolar switched current mirror driver.
Bipolar drivers are usually terminated into a midrail termination voltage. The result is a bipolar drive current. A pair of current mirrors is used to generate the bias voltages. Similar methods can be 6 With unipolar differential signaling. No reference is required. In this case the two logic levels are denoted by equal amounts of current flowing in opposite directions. Waveforms for the rise-time-controlled driver are shown at the right side ofthe figure.. To avoid this.
Like Goldilocks's porridge. On the other hand. The driver is divided into four current-mode drivers. Depending on the degree of accuracy needed in the rise time. As a result.. If the current drivers turned on instantaneously. The segments of the driver are sequenced on by the taps of a delay line so that each segment turns on 0.
Too fast a rise time couples energy into parasitic tank circuits Section 6. Because of the gain of the output stage. Because the rise time of a typical CMOS gate can vary by 2: The RC output of the predriver gives a long predriver rise time.
Until the slowly rising predriver output reaches the threshold voltage of the output stage. The figure shows an ideal predriver with an RC output circuit driving a switched current-source output driver. Waveforms of the circuit are shown at the right side of the figure.
The situation is illustrated in Figure Consider the abstract voltage-mode driver circuit shown in Figure Because of the delay. Once the predriver voltage reaches the output threshold. This results in a relatively fast output rise time. To get a tr of half a bit cell. RC rise-time control is not a viable technique for rise times that are greater than about one-tenth of the bit cell. This results in a long output delay. On a falling output. When the driver switches from high to low.
The pull-up and pull-down drivers are segmented into four smaller drivers with equal output impedance. Suppose there are n stages. When the input switches from low to high. Table shows the values of conductance and resistance for a four-stage voltage-mode driver with a [2 termination. Only three stages are shown because the fourth stage.
Each driver is illustrated here as a pair of resistors selectively connected to the output by a switch. A reference signal. As shown in the waveforms at the right side of the figure.
To reject noise primarily due to IR supply variations. When the line is not discharged. Figure shows the latter approach. At each intermediate point during the switching process the connected resistors form a voltage divider so that the switching linearly ramps the open-circuit output voltage between the two supplies..
To see that the ramp is linear. The voltage swing on the line is set by the width of the discharge pulse. Adaptations of the drivers described above for terminated transmission lines can achieve both of these functions.
Care is required in routing the reference signal to keep its capacitance balanced with that of the line and to make most noise couple equally into the line and the reference so it can be canceled.
This pulse is usually generated by a circuit that tracks the characteristics of the driver and the line so that the voltage swing remains constant across process. The driver gives a linear ramp on in current and the ramp off is usually controlled by the driver devices dropping out of saturation as the voltage approaches the supply. Its output impedance damps the tank circuit. This gives the circuit a break-before-make action that eliminates overlap current.
The transmitter multiplexes a four-bit input. It is useful to think of the earlier of these two clocks as the on-clock because it turns the current source on and the later clock as the off-clock because it turns the current source off. Each input is tied to one gate of a stack of three NFETs that form a clocked current source tied to the line. The other two gates are tied to two clocks. A simple multiplexing transmitter is illustrated in Figure The current source drives the line only when both clocks are high.
A transmitter with gain less than 1 can have bandwidth greater than the gain-bandwidth product of the technology by multiplexing several output signals together in the transmitter under control of a multiphase clock.
When operated as bipolar. Stated differently. In some cases feedback from the line or reference is used to terminate the pulse once the line has been discharged to the desired voltage. When driving LC loads. This circuit is the parallel combination of two stacks with the transistor positions reversed between the two stacks. Better performance and reduced area can be achieved by qualifying the off-clock with the data and driving a stack of two FETs with the on-clock and this qualified off-clock as illustrated in Figure The circuit of Figure uses a stack of three FETs to combine the two clocks with the data signal.
The result is that the two input clocks see the same effective threshold and thus switch without relative phase shift. The off clock. Because the on-clock and off-clock drive transistors at different positions in the stack.
The figure shows one multiplexer input. This effect can be eliminated by replacing the clocked part of each stack in Figure with a two-input symmetric gate. The circuit of Figure introduces timing offsets because the transistors at different positions in the stack have different threshold voltages due to the body effect and thus switch at different points in time. The multiplexer must be fast enough for its output. The input multiplexer accepts full-swing inputs dO: The circuit of Figure consists of three stages: A multiplexed transmitter can also be implemented by multiplexing before the predriver and output stage.
Each input of the multiplexer consists of a series pair of NFET pass gates controlled by an on-clock and an off-clock. This arrangement requires less circuitry and puts a smaller capacitive load on the output than the output-multiplexed circuit of Figure Voo eliminates dead time in the operation of the output stage. The figure shows a prototypical data eye Chapter 6 in [LeeMess94] representing the physical quantity encoding one binary symbol.
Systematic jitter occurs when the duration of the output bits is uneven in a repeatable manner. This systematic error can be a factor in an overall timing budget. The resolution and offset in these two dimensions determine the size of the eyeopening required for reliable detection of the signal.
Shortly after the predriver starts switching. The light shaded rectangle indicates the size of the gross timing margin width and the gross voltage margin height. In the voltage dimension. For proper operation. In the time dimension. In this section we will be concerned with receivers that detect voltage. A multiplexing transmitter may exhibit systematic jitter if the phases of the multi phase clock used to control the multiplexer are not evenly distributed or if delays between the clock signals and the output are not carefully matched.
A receiver should have good resolution and low offset in both the voltage and time dimensions. The dark-shaded rectangle represents the aperture time width and sensitivity height of the receiver.
The second advantage is reduced timing uncertainty.
For these reasons. A regenerative clocked amplifier can amplify the smallest signals up to full swing. There are four main advantages to combining the functions of detection and sampling into a single circuit. The separate design adds two elements to the timing budget: A bank of clocked amplifiers operating as a demultiplexing receiver Section Only if the dark rectangle remains entirely within the light rectangle.
For completeness. As shown in Figure b. Only during the aperture time of the flip-flop is the detected value. Figure a shows the two tasks separated. A static amplifier. With the single-clocked amplifier. The signal x is then sampled by an edge-triggered flip-flop with a clock. By choosing one or both offsets to be negative.
With the functions separated. A reference inverter. This technique in effect turns the inverter into a sequential differential amplifier with the primary source of offset becoming the variation in the reference over time. Inverters can also be used effectively by compensating for their variations. Combining the sensitivity with the offset.
One approach is to use a switched capacitor circuit to measure and cancel the offset voltage of the inverter. A gain ofthis magnitude usually requires stages of amplification. For a tight noise budget. This analysis should not imply that the inverter is without merit for use in a static receiver. More importantly. For some processes with lower DC gains. There are two ways inverters can be used effectively in receivers. As long as the threshold voltage of the receiver inverter tracks the threshold voltage of the reference inverter.
For this reason inverters are in widespread use as receivers on CMOS chips. Recall from Section 4. The input-referenced offset voltage of the inverter is reduced by the gain of the input stages and hence is less of a problem in this configuration. Multiplying the inverter gain 20 by the gain of the earlier stages gives a high overall gain and hence a sensitive amplifier. Another approach to compensating for the variation in the threshold voltage of inverter receivers is illustrated in Figure In - IAI With an output swing of 2.
If the current bias is too low. The offset voltage for these circuits depends primarily on the matching of two identical components e. One approach to biasing a source-coupled amplifier is to use a replica-biased circuit.
Static differential amplifiers can be built using either a source-coupled pair Section 4. Their primary advantage is a relatively low offset voltage compared with an inverter. An alternative approach is to use a self-biasing amplifier such as the Chappell amplifier [Chap88] shown in Figure The current-source bias must be set to balance output voltage swing against input common-mode range. Static differential amplifiers also are largely insensitive to power supply variation in contrast to the inverter.
Most source-coupled differential amplifiers have a limited common-mode operating range. At the same time. The proper bias depends on process variation.
This circuit sets the bias voltage to give the desired output swing into a given load circuit. An excessive bias. The input voltage must be high enough to bias the current source on and put the input transistors into the saturation region. The single-ended Chappell amplifier Figure a is a source-coupled Jut out in-. This results in negative feedback that drives the bias voltage to the proper operating point. A symmetric variation on the Chappell design that we have found useful is shown in Figure b.
This circuit uses two current mirrors for the load. The selection depends on the requirements for common-mode range and input isolation. The gate-isolated sense-amplifier of Figure c provides complete input isolation but at the cost of limiting the input common-mode range. The complementary clocked sense-amplifier of Figure b has the advantage of a rail-to-rail common-mode range.
Either of the clocked differential amplifiers described in Section 4. The negative feedback control of the bias current is the same as for the straight Chappel amplifier. The resulting amplifier has a large common-mode range. One approach to increasing the input dynamic range of a source-coupled amplifier is to combine an NFET source-coupled pair with a PFET source coupled-pair. If the bias current is too low high.
The single-ended Chappell amplifier is not suitable for driving subsequent stages of differential amplifiers.
Dally W.J., Poulton J.W. Digital Systems Engineering
Figure shows how this is done in two steps: When the clock. This gives good rejection of timing noise and low-frequency compared with the data rate voltage noise. We can describe the operation of the amplifier as sensing the convolution of the impulse response. Because this amplifier is integrating only half the time and precharging the other half. This has the advantage of rejecting highfrequency noise that will tend to average out over a bit cell but has the disadvantage of reducing immunity to low-frequency noise because the low-frequency noise is being integrated across times where the signal amplitude is reduced.
In the presence of high-frequency noise of significant amplitude. At the end of the integration period. The sense node lags the input because of the RC delay of the transmission gate. As shown in Figure c. This gives a reversed impulse response that exponentially falls off to the left with the RC time constant of the transmission gate.
These are all shown time reversed h -t so that they correspond to the weight applied to the input signal at a given point in time. It responds with equal weight to the input signal during the entire integration period. It ramps up with the rise time of the clock to a peak value and then falls off exponentially with the time constant of the regenerative circuit as the sense nodes of the amplifier diverge.
One method for shaping the current pulse is. Optimal filtering theory tells us that the ideal impulse response is that of a matched filter. In the presence of timing noise. The ramp from the peak of the response to zero is set by the rise-fall times of the clocks controlling the transmission gates. A filter matched to the impulse response of the data has a timereversed impulse response that has a shape identical to that of the data eye.
When the clock rises. This response is almost exactly the reverse of the pass-gate input sense amplifier. Figure b shows the impulse response of a gate-isolated sense amplifier Figure c. A complementary clocked sense amplifier Figure b has the impulse response shown in Figure a. For the first two waveforms.
The response is zero until the clock rises. As illustrated in Figure A clock with n equally spaced phases sequences an array of n sense amplifiers to take n equally spaced samples of an input signal each clock cycle. Here the gate voltage for the current-source transistor. In practice. With the appropriate choice of parameters. The current pulse is foreshortened. The higher rate is required on the receiver to recover timing information as well as data.
The eight outputs of the receive amplifiers. S Tracking receivers. With a multiplexing receiver. Oversampling clock receivers require at least three samples per bit cell.
To stretch the period during which all eight signals are valid. Figure shows a 1: The waveforms show the situation where the even phases are used to sample cell edges for clock recovery and the odd phases are used to sample the data. Waveforms illustrating typical operation ofthis circuit are shown in Figure A single input signal. This stretches the valid period to five phases. These receivers typically recover timing information by periodically shifting their sample points by a half bit cell to sample the timing and then shifting the sampling back to resume data sampling.
An adjustable highvoltage power supply charges a capacitor to a preset voltage. A typical current waveform is shown in Figure c. A lumped circuit equivalent for an HBM tester.
This is particularly true in dry climates and air-conditioned environments in which a test-assembly technician can easily acquire a static body charge.
Digital system engineering
Many Ie fabricators use an instrument that simulates this type of event. The circuits on a chip that are connected to the chip's external interface the bond pads for input. The goal of electrostatic discharge ESD protection circuitry is to allow such an event to occur without damage to the chip's internals. These high-voltage stressing events almost always occur when a chip is being handled by a person or a robot during test.. The ESD protection circuitry in the chip clamps the voltage between any pair of pins to a few volts.
A spark. Rise time is usually very fast compared with the event duration. Other types of testers are. For a typical ESD test. As the technician's hand closely approaches a chip. Ensuring higher levels of ESD protection therefore inevitably compromises performance. FET gates can also be stressed in a secondary way by large fields between source and drain.
Class I devices are allowed to fail at 2 kV. Class 2 devices fail between 2 and 3 kV. The result is a permanent shift in the device's threshold voltage and consequent degradation of performance. Tox is 70 X During normal chip operation these devices are never energized and simply represent low-pass filters between pad and internal circuitry.
Class 3 devices must withstand up to 4 kV. Because an ESD event can drive current into a chip between any pair of pins. A particularly unpleasant scenario is an ESD event that occurs when a chip is powered up. ESD protection circuitry uses large-area parasitic devices of various kinds attached to the bond pads and series impedances between pads and internal circuitry.
In our example 0. Although ESD damage often occurs during handling and assembly operations. The DoD standard defines various classes ofESD protection according to failure voltage threshold.
Under these circumstances. After the series of stresses has been applied. The critical voltage for breakdown in Si02 is about 7 x VIM. As fabrication technology scales down. When the drain current reaches a critical value. Neither of these breakdown processes is necessarily fatal to the parasitic diode.
The interconnect wiring. This positive feedback process. This breakdown phenomenon is not necessarily destructive aside from the likelihood of hot-electron wear-out. During ESD events. The voltage. The multiplication of carriers is called avalanche breakdown and leads to rapidly increasing conduction. A plot of breakdown drain current for an FET is shown in Figure If the junction is a FET source-drain terminal.
The source and drain diffusion terminals of an FET form diodes with their associated substrate material. Under reverse bias. I CRIT. This phenomenon leads to a third breakdown mechanism. At high reverse voltages. These diodes are normally lightly reverse biased and are considered parasitic elements during normal chip operation.
The hole-electron pairs that are generated themselves become carriers. Field-effect transistors can be thermally damaged by more complex mechanisms. Electrostatic discharge currents. Even in the absence of carriers with sufficient energy to ionize lattice atoms. The critical current feri! An ESD protection network is usually composed as shown in Figure and consists of four elements.
Because ESD currents are quite large. It must handle very large currents. The primary shunt devices are the first line of defense against ESD damage. The secondary shunt is referred to the local power supply of the protected elements to minimize the voltage across individual devices.
The power supply networks are the most robustly wired nodes in a chip.
The secondary shunt further limits the voltage excursion presented to the protected circuitry. The current that flows through these shunts is usually quite small and is limited by the series element.
The series impedance. Current from positive ESD events is carried by normal FET drain current for low-voltage events and by punchthrough for larger ones. Such field devices are usually considered unwanted parasitics. During a positive-going ESD event. The ESD ground network carries the current between the two pads through the complementary devices in each pad.
Note that this strategy works even if the ESD current flows into one signal pad and out another. The punchthrough phenomenon is enhanced by the presence of a gate terminal. During negative-going ESD events. Both Voo and ground supply networks are co-opted to carry ESD currents away from these devices. The leading edge of the event forward biases the BE junction of the PNP protection device in the upper pad. By examining the cross section of the dual-diode arrangement.
A typical arrangement of layers is shown in Figure b. These bipolar devices are critical to the correct operation of the dual-diode primary shunt for signal-pad-to-signal-pad ESD events. The clamp voltage for these junction devices is essentially independent of other CMOS process variables. Because latchup during circuit operation is usually fatal to the chip.
This device also called a silicon-controlled rectifier rSCR] or thyristor is usually undesirable. A combination of. During normal power-up. The inverter's input is an RC delay circuit with a time constant of 10 itS or so. With both devices on. This shunt current encourages ESD currents to flow mainly in the BE junctions of the protection devices.
To avoid depending on the purposely poor bipolar emitter-collector conduction to carry ESD currents.
The shunt contains a very wide. A power supply shunt. Some IC manufacturers provide more exotic primary shunt devices. A very large resistance Ro discharges the capacitor during power-down.
Heat generated in the resistor is not easily shed and could lead to failure of the device. In older CMOS processes. If the protected node tries to swing below ground. Although diffusion resistors are possible.
Usually the series resistor is of the order of a few hundred ohms. A typical circuit is shown in Figure Some processes offer a "silicide blank" mask layer that allows the designer to remove the silicide on poly and diffusion in selected areas.
N-well resistors are a good option in processes in which the N-well resistance is well controlled and characterized. Input capacitance for a typical receiver is of the order of Older designs sometimes used polysilicon resistors. If the signal tries to swing above the local Voo voltage.
A better option is to place the resistor in the silicon substrate. Poly without silicide has resistance of a few tens of ohms per square. Poly is also completely surrounded by a thick field oxide whose thermal conductivity is low 1. This feature allows resistors to be built using poly. Polysilicon resistors are not a particularly good idea in new designs.
During an ESD event. If the burst of electrons produced during such an event gets collected on the N-well of a normal CMOS circuit within the chip.
These measures include the addition of small-value resistors in series with the drain of each FET connected to a pad. This can result in overheating nearby metal-to-silicon contacts. ESD primary shunts and the FETs in pad output drivers are spaced some distance away from chip internal circuitry. Special layout rules are usually required for FETs that drive pads. It is often advantageous to place the primary shunt devices out at the die perimeter.
As previously mentioned. These carriers can propagate for some distance. These and other measures tend to reduce the speed of output drivers. They may.
Most fabricators require the addition of primary shunt diodes on outputs. Future signaling system designs will not be able to take full advantage of device scaling because of these problems.
In modem processes and especially for low-swing. In some cases. In the case of a below-ground signal excursion. The extra distance between emitter and possible. Guard rings are always heavily contacted and robustly wired to the power supply networks in continuous metal. Ensuring uniform current distribution is especially important in FETs whose drains are connected to a pad. This section is intended merely as an introduction to a complex topic.
Care is required in the layout of conductors and via-contact arrays that carry these currents. The arrangement shown in Figure ll b should be avoided.
The illustration shows a metal-to-silicon connection typical of an ESD protection diode. Sources and drains for driver FETs should be contacted uniformly and as densely as possible along the full width of the device.
Failure to do so could lead to current crowding in the FET channel. The written guides for this part of the design process tend to be hard to obtain from vendors and difficult to understand. In Figure ll a. This arrangement leads to a uniform current distribution across the contacts in the horizontal direction.
IC fabrication vendors differ substantially in their recommendations for the construction of effective ESD protection devices. Current is delivered from one side of the contact array. These are very effective collectors of wandering minority carriers. In particular. Some of these considerations are outlined in Figure A single. Three termination schemes transmitter-only or self-source-terminated. An automatically adjusted bias control sets the signal current level.
The TxClk is converted to differential form if necessary and driven into a delay-locked loop. The transmitter accepts 4 bits of data from the chip internals each clock cycle and transmits them over a differential pair at 1 Gbitls.
The loop is locked around four delay stages in such a way that the phase shift between output clocks cO and c3 is The DLL is implemented with source-coupled.
Terminators are built exclusively with PFETs and are automatically adjusted using a thermometer code by a controller that uses an off-chip reference resistor. A multiphase clock generator using a delay-locked loop DLL controls highspeed operation of the transmitter. Edge-rate control is employed in the transmitter to limit rise and fall times on the output to about half a bit-cell time. The example subsystem will be composed entirely of point-to-point links that use current-mode differential signaling.
It uses VDD-referenced current mode signaling with source-coupled current steering.
This termination scheme was described in some detail in Sections Transmitter and receiver chip internals are assumed to be clocked at MHz. Autoadjusted on-chip terminators are used. We will assume that. The driver is segmented into four equal-sized pieces for edge-rate control. Finer control is needed.. The even-numbered clocks are on I-ns boundaries and become the bit-cell clocks for the transmitter. Design considerations for the output driver are determined partly by the ESD rules for the process and by related issues of layout convenience..
A 5-ma signaling current will be assumed These closely spaced clocks are used to drive a segmented transmitter. Timing of the clocks is outlined in Figure The circuits needed to implement the multiphase clock generator are discussed in Chapter 12 in some detail.
We will assume that these Q resistors are made from N-well and that their maximum resistance under process and temperature variation is Q. A total transmitter current of 5 rna gives m V of signal mV differential if only one termination is used.
One possible circuit implementation is shown in Figure Because the tail current is not critical. In either case the signal is about an order of magnitude larger than the fixed sources of noise we estimated for this type of signaling system in Section 6.
This voltage is dropped across a resistive voltage divider conveniently built using N-well resistors or perhaps triode-connected PFETs. The physical basis for the reference is the threshold voltage of the PFET. On the right-hand side of the figure is a replica transmitter scaled down to one-quarter. Figure shows the details of a possible layout that is particularly convenient.
As a first cut. A "poor man's" voltage reference. It remains to choose a suitable value for W. The tail current sets the maximum signal voltage swing across the terminators. The current tail transistor must be well into saturation at the desired signaling current. One of many possible ways to generate the tail bias is to produce a reference voltage equal to the intended signaling voltage and then servo the tail bias voltage in a replica transmitter circuit.
The tail current is servoed to maintain this voltage. The outputs should switch as a complementary pair. All other FETs are minimum length. The ground return for the replica transmitter should be the same as that for the driver segments.
Figure shows an implementation of a predriver using the methods described in Section The voltage crossover between rising and falling outputs of the predriver holds both steering devices on at the midpoint of the switching event. An autoadjusted termination resistor is included. The signal swing out of the pre driver is minimized. J Autoadjusted. To obtain this behavior. The rising predriver output almost immediately begins turning on its steering transistor. The voltage swing on dH.
This requires a make-before-break action in the current-steering transistors to avoid allowing the current tail transistor to fall out of saturation and the tail voltage to collapse. VB is bypassed using an NMOS capacitor to the transmitter-local ground so that the bias voltage will ride up and down with any ground noise that is generated. A real design would have to consider loop stability.
This choice for VLOW out of the predriver optimizes several aspects of driver behavior. Lm length to help track the behavior of the output driver.
Note that the current tails are all drawn at O. The package is assumed to have an internal ground plane with no additional inductance between package ground and board ground. The transmitter's differential output pads are interspersed with ground pins pin order: Figure outlines the package model. The Off clock for each bit is the complement of the On clock for the next data bit.
Two l-pF capacitors model the bond pads and ESD diodes. Each data bit is gated into the predriver by an "On" clock.
As for the bond wires. Signals dO and d 1 are sent during the first half of the TxClk cycle. The package's internal signal traces are assumed to be 3-mil-wide. Figure a shows the differential signal seen at the receiver when modeled with a lossless line and no package parasitics. The layer stackup assumed in this model is shown in Figure The transmitter was simulated using a pseudorandom bit pattern for about 80 bit-cell times.
Nominal fabrication models. Identical package models are assumed for both transmitter and receiver.. Figure b shows the effects of the package parasitics. The transmitter's clocks are driven by voltage sources rather than the DLL.
Lossy model. This model is for reference purposes only. A small amount of overshoot is evident owing to slight mismatch at the PFET terminators.
Frequency-dependent attenuation is included in the model. Lossy-U model with two connector interfaces e. Connectors are handled with a vendor-supplied model for a standard 2-mm backplane connector. Figure ll d shows the additional effects of a pair of connectors inserted into the line. SNR has been degraded to about 7: Figure shows eye diagrams for each of these cases.
SNR has been reduced to about Figure c shows the effects of the lossy transmission line model. I versus 5: I as for either of the single-termination schemes.. The magnitude of the signal at the receiver is twice as large for both transmitter-only and receiver-only termination schemes. The frequency-dependent attenuation has introduced both noise and timing jitter. It is also necessary to include power supply and substrate voltage fluctuations.
Fixed noise sources include line attenuation already included in the transmission line model and receiver offset. Improvement would also be larger for a best-case simulation. For the example system. Jitter at the receiver arises from frequency-dependent line attenuation. Choice of termination scheme would. Differential coupling coefficients are generally quite low one of the many benefits of differential signaling.
Figure shows eye diagrams before and after addition of all of the above noise sources.
Both simulations use the transmission line model with lossy line and connectors. Edge-rate control might be expected to yield greater benefit in situations where package and connector parasitics are larger than in this example signaling system.
Proportional noise sources include cross talk from neighboring comparable signals and noise induced by continuous adjustment of terminators as well as the intersymbol interference from line reflections already handled by the line and package models. Cross talk is modeled using a second neighboring transmission line carrying a signal of twice the amplitude lO-ma signal current to model the effects of two neighboring lines and a series of.
A mV offset is introduced at the receiver. For the simulation in the figure. Whether hand-generated or commercial interconnect and packaging models are used. It is relatively easy to build receiver sense amplifiers with apertures of 50 ps or so in the example technology. Design of a "real" high-performance signaling system requires much greater attention to the details of the transmission line system than outlined in this example.
Most of the additional noise in Figure 1I b results from power supply noise. In a "real" design. The eye diagram has an apparent SNR of about 3: A block diagram ofthe receiver is shown in Figure With the simulated noise sources we have used here. In some designs. In this example. Variation in the PC board trace impedance over manufacturing tolerances should also be included. A control clock. The phase shifter is built using a DLL similar to the transmitter's clock generator and is locked so that four delay stages generate half of a 4-ns clock period of delay.
A set of retiming latches brings the data samples into the RxClk domain. It uses the same control voltage as the DLL and is built with identical delay elements. Methods for Addition and Subtraction. Fast adder architectures. Multi-operand addition. High-throughput arithmetic. Low-power arithmetic.
The course uses the ARM processor core as an exemplar of a modern processor architecture that is now ubiquitous in embedded systems. This material will include: Variables, data-types and arithmetic expressions. Strings, Loops, Arrays. Functions, Structures, Pointers, bit operators. The pre-processor. Debugging Programs. Object-Oriented Programming. The Standard C Library. Issues such as software testing and testing strategies are discussed. GNU based toolchains for Microcontroller development.
Network security and encryption mechanisms: IPSec and other security protocols. Network performance analysis, queuing theory, and network simulation. Practical work is carried out in air-conditioned laboratories, with state-of-the-art equipment and outstanding IT infrastructure. Stage 1 modules are assessed by coursework and examination at the end of the year.
All years include project work to replicate industrial practice and develop skills to maximise employability. Please note that progression thresholds apply. The precise breakdown of hours will be subject dependent and will vary according to modules. Please refer to the individual module details under Course Structure.
Methods of assessment will vary according to subject specialism and individual modules. Programme aims The programme aims to: educate students to become engineers, well-equipped for professional careers in development, research and production in industry and universities, and capable of meeting the challenges of a rapidly changing subject produce computer systems engineers with specialist skills in hardware and software engineering, prepared for the complexities of modern computer system design enable students to satisfy the professional requirements of the IET provide academic guidance and welfare support for all students create an atmosphere of co-operation and partnership between staff and students, and offer students an environment where they can develop their potential produce high-calibre, professional engineers with advanced knowledge of modern embedded electronic systems enable students to fully satisfy all of the educational requirements for Membership of the IET and Chartered Engineer status.
You gain transferable skills including: the ability to generate, analyse, present and interpret data the use of information and communications technology personal and interpersonal skills and working as a member of a team effective communication in writing, verbally and through drawings effective learning for the purpose of continuing professional development critical thinking, reasoning and reflection how to manage time and resources within an individual project and a group project.Proposal engineering Proposal engineering is the application of scientific and mathematical principles to design, construct, and operate a cost-effective proposal development system.
A discussion of clock distribution techniques closes the chapter. To define the signal value unambiguously during a particular clock cycle. As technology continues to advance. This results in a relatively fast output rise time. The package is assumed to have an internal ground plane with no additional inductance between package ground and board ground.
A second delay line gets the output of the phase shifter and generates the eight sample clocks. This technique in effect turns the inverter into a sequential differential amplifier with the primary source of offset becoming the variation in the reference over time. Frequency-dependent attenuation is included in the model.
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